The transmission of radio signals carrying data in modern wireless communications can be realized based on a number of different communications systems, often specified by a standard. Mobile radio receiver devices include analog radio frequency (RF)/intermediate frequency (IF) stages which are arranged to receive and transmit wireless signals via one or more antennas. The output of the RF/IF stage is typically converted to baseband, wherein an analog to digital converter (ADC) converts incoming analog signals to digital samples, which are then processed for signal detection and decoding of the data in the form of reliability values. The ADC may alternatively operate directly at IF, in which case the conversion to baseband is performed in the digital domain.
In a wideband CDMA (wideband code division multiple access) cellular system, different physical channels are multiplexed in the code domain using separate spreading sequences called orthogonal variable spreading factor (OVSF) codes. In a case where multiple transmit antennas are used, the same spreading and scrambling codes modulate symbols transmitted from both antennas, but the symbol sequence is different. FIG. 1 is a schematic diagram illustrating two transmit antennas A1, A2. For each antenna A1, A2, a symbol sequence is supplied to a multiplier M1, M2 which multiplies a symbol sequence with a spreading/scrambling code. The transmit antennas A1, A2 are intended to transmit on the same channelization code (in the case of FIG. 1 the CPICH (common pilot channel) downlink code). An adder ADD1, ADD2 allows other channels to be added into the transmission from each of the antennas.
FIG. 1 schematically illustrates dual antenna transmission in the case where the CPICH channel comprises a plurality of pilot symbols, with the data being transmitted using the open-loop transmit diversity (space-time transmit diversity (STTD)) scheme specified by the 3GPP WCDMA system (see, e.g., the 3GPP specification TS 25.211, “Technical Specification Group Radio Access Network; Physical Channels and Mapping of Transport Channels onto Physical Channels (FDD)”, December 2005), or the closed-loop transmit diversity (transmit adaptive array (TxAA)) scheme specified by the 3GGP WCDMA system (see, e.g., the 3GPP specification TS 25.214, “Technical Specification Group Radio Access Network; Physical Layer Procedures (FDD)”, June 2005). The use of multiple transmit antennas requires the estimation of the channel from each transmit antenna at the receiver in the user equipment (UE).
As shown in FIG. 1, for support of channel estimation at the user equipment (UE) receiver, a different symbol sequence is transmitted from each antenna. The modulated CPICH symbol sequence for each transmit antenna is shown in FIG. 2. The symbol S in FIG. 2 is constant, S=(1+j)/√{square root over (2)}. The CPICH spreading factor is 256 and the same spreading and scrambling codes modulate the symbols for both antennas. Antenna A1 transmits the symbol d1(k) always equal to S,d1(k)=S, where antenna A2 transmits the symbol d2(k) equal to +S or −S,d2(k)=ξ(k)·S whereξ(k)=(−1)└(k+1)/2┘,and k is the symbol index counted from the CPICH frame boundary.
As can be seen from FIG. 2, the sign verifies the following property,ξ(2k)+ξ(2k+1)=(−1)k+(−1)k+1=0.
FIG. 3 shows the descrambling/despreading circuitry of the CPICH at the UE, for different values of delay applied to the received signal samples corresponding to the taps of the channel impulse response. The circuitry comprises a set of fingers indicated by the reference numeral 2, each for descrambling a delayed version of the received signal. The signal at the output of the CPICH descrambling/despreading corresponding to the l-th delay of the channel, l=1, . . . , L, isy(l,k)=Sh1(l,k)+ξ(k)Sh2(l,k)+n(l,k),where h1(l,k) (respectively h2(l,k)) is the channel gain from antenna A1 (respectively antenna A2) corresponding to the l-th channel delay, and k is the symbol index.
The channel estimation is performed in the same way for each value of delay. For application to the conventional rake receiver processing, the selection of the delays for which the channel estimation is performed is done in a way to capture most of the channel energy. The exact implementation of the delay selection is out of the scope of the present description and is known to a person skilled in the art.
For simplicity, in the following we will omit the delay index, so that the signal after descrambling/despreading is written asy(k)=Sh1(k)+ξ(k)Sh2(k)+n(k).
The estimation of the l-th channel tap is performed separately for the channel from each antenna, h1(k) and h2(k).
The problem then is how to exploit best the received pilot symbols for the channel estimation.
We introduce the variables:
                    z        1            ⁡              (        k        )              =                                        y            ⁡                          (                              2                ⁢                k                            )                                +                      y            ⁡                          (                                                2                  ⁢                  k                                +                1                            )                                      2            ·              S        *                                          z          2                ⁡                  (          k          )                    =                                                                  ξ                ⁡                                  (                                      2                    ⁢                    k                                    )                                            ⁢                              y                ⁡                                  (                                      2                    ⁢                    k                                    )                                                      +                                          ξ                ⁡                                  (                                                            2                      ⁢                      k                                        +                    1                                    )                                            ⁢                              y                ⁡                                  (                                                            2                      ⁢                      k                                        +                    1                                    )                                                              2                ·                  S          *                      ,  where asterisk denotes complex conjugate.
For a slowly varying channel h1(2k+1)≈h1(2k) and h2(2k+1)≈h2(2k), and due to the property ξ(2k)+ξ(2k+1)=0
            z      1        ⁡          (      k      )        ≈                    h        1            ⁡              (                  2          ⁢          k                )              +                                        n            ⁡                          (                              2                ⁢                k                            )                                +                      n            ⁡                          (                                                2                  ⁢                  k                                +                1                            )                                      2            ·                        S          *                .                                  ⁢                              z            2                    ⁡                      (            k            )                                ≈                    h        2            ⁡              (                  2          ⁢          k                )              +                                                      ξ              ⁡                              (                                  2                  ⁢                  k                                )                                      ⁢                          n              ⁡                              (                                  2                  ⁢                  k                                )                                              +                                    ξ              ⁡                              (                                                      2                    ⁢                    k                                    +                  1                                )                                      ⁢                          n              ⁡                              (                                                      2                    ⁢                    k                                    +                  1                                )                                                    2            ·                        S          *                .            
Therefore, for a slowly varying channel, z1(k) (respectively z2(k)) is a noisy estimate for the channel h1(2k) (respectively h2(2k)).
One approach (e.g., as cited in U.S. patent application Ser. No. 10/139,904, “Transmit Diversity Pilot Processing”, published November 2003) exploits the above property. The channel estimate is performed once every two CPICH symbolsh1(2k+1)≈h1(2k)=f(z1(k+L1), . . . ,z1(k+1),z1(k),z1(k−1), . . . ,z1(k−L2))h2(2k+1)≈h2(2k)=f(z2(k+L1), . . . ,z2(k+1),z2(k),z2(k−1), . . . ,z2(k−L2)),where (•) is a filtering function, with L1 and L2 the length of the anti-causal and causal parts of the filter response, respectively. For an infinite impulse response (IIR) filter, the length of the causal part is infinite, L2=+∞.
For highly time-varying channels, the approximation h1(2k+1)≈h1(2k) and h2(2k+1)≈h2(2k) are no more valid, and the above approach causes a degradation of the quality of the estimated channel.
Another method based on the estimation of the sum and the difference of the channels of the two antennas A1 and A2, is proposed in U.S. patent application Ser. No. 10/139,904, “Transmit Diversity Pilot Processing”, published 6 Nov. 2003. However, the method proposed in this patent does not update the channel estimate for each antenna every CPICH symbol, and leads to an increased complexity if a finite impulse response (FIR) filter is used to improve the channel estimation. As the pattern of the received signal of the sum and the difference of the channels from the two antennas is not uniform, this requires the use of a different FIR filter depending on the position.